Frequency offset differential pulse position modulation

ABSTRACT

The present invention provides such a need by utilizing a frequency offset differential pulse position modulation scheme to transmit data between computing devices within a wireless network system. The differential pulse position modulation component of the scheme enables the present invention to provide relative immunity to interference for the system. In particular, such immunity from interference is achieved by utilizing a blanking time between pulse positions, which is large enough to allow the interference between frequency offset—differential pulse position modulation pulses to subside. The frequency offset component of the scheme enables the system to utilize multiple frequency channels to enable the system to achieve higher data rates. In particular, by utilizing a time offset between the frequency channels, the blanking time can be reduced, thereby increasing the amount of data that can be transmitted with a set period of time.

RELATED APPLICATION

This application claims the benefit under 35 U.S.C. §119(e) of U.S.Provisional Application Serial No. 60/119,225 filed Feb. 8, 1999,entitled “Frequency Offset Differential Pulse Position ModulationScheme,” having common inventorship and assignee, which is incorporatedby reference in its entirety herein.

BACKGROUND OF INVENTION

1. Field of Technology

The present invention generally relates to computer networks and, moreparticularly, to wireless networking.

2. Description of Related Art

Throughout the 1980s, local area networks (“LANs”) emerged as a newtechnology market. Aided by technology that allowed LANs to operate overthe structured unshielded twisted pair telephone wire already installedin most buildings, the networking market has become a $7 billion a yearbusiness. Unfortunately, as organizations grow more dependent on thesehigh speed LANs, structured wiring systems have become less palatable.Current LANs are highly dynamic, marked by constant moves, additions andconfiguration changes to keep the system operating at peak performance.As a result, operating and maintenance costs are growing as usersattempt to keep pace with this rapidly evolving environment. Typically,more than 60 percent of conventional LAN reconfiguration costs areattributable to labor, with the percentage much higher in metropolitanareas where labor is more expensive. For example, a conventional highspeed LAN (e.g. a 30 node LAN) typically requires an average of threeweeks to plan and install. In addition, more than 35 percent of problemswith LANs using hubs and structured wiring and 70 percent of problemswith LANs without hubs and structured wiring typically are attributableto the cabling of the LAN. In particular, the costs to move an averageEthernet unshielded twisted pair (“UTP”) connection of such a LANaverage approximately $500 or more (in parts and labor) per node. Thiscost further increases in major metropolitan areas where increased laborcosts and a prevalence of fiber-based LANs exist.

In the early 1990s, an alternative to wired LANs was the wireless LAN(“WLAN”). The benefits of this wireless communication technique in suchan environment relates to the avoidance of much of the costs associatedwith the cabling of the network. However, most of these WLAN productsfailed to meet several key criteria essential to wide-scale adoption.The system failed to be fast (e.g. at least 10 Megabits per second), wasnot simple (e.g. plug and play) and was not economical (e.g. less than$500 per node).

In a manner analogous to the growth of the wired LANs, initialapplication and market success of the WLAN was in specialized, verticalmarkets. Thus, applications that highly valued the mobile, untetheredconnectivity were the early targets of the WLAN industry. Over the lasttwo years, however, WLANs began to emerge as an option to fill anever-widening gap in the corporate enterprise infrastructure. Currently,there are three basic types of wireless networks: wide-area network(“WAN”), local area networks (“LAN”) and campus area networks (“CAN”).

These WLANs, however, exhibit one drawback or another such as being tooslow, restricted to certain environments or too expensive. In an idealenvironment, the optimal WLAN solution would eliminate theseshortcomings and exhibit the characteristics of being easy to install,having performance of over 10 Megabits per second (“Mbps”), being costeffective and being compatible with existing Ethernet networkingequipment. Unfortunately conventional WLANs do not meet more than a fewof these requirements with some failing to satisfy even one.

Today's fourth generation (“4G”) of WLAN technology attempts to resolvethese problems as well as increase bandwidth and avoid the interferenceassociated with the more crowded lower frequency bands by operatingwithin the 5 GHz frequency band. This increase in frequency enables such4G WLANs to operate at data rates of approximately 10 Mbps. Such WLANstypically operate in the 5.775-5.850 GHz Industrial, Scientific, andMedical (“ISM”) band as well as within frequency bands at 5.2 GHz and5.3 GHz, Unlicensed-National Information Infrastructure (U-NII) bands.These three segments have been designated by the FCC as exclusively forhigh speed data transmission. Unlike the lower frequency bands used inprior generations of WLANs, the 5 GHz bands do not have a large numberof potential interferors, such as microwave ovens or industrial heatingsystems. In addition, there is much more bandwidth available at 5GHz-350 MHz as compared to the 83 MHz within the 2.4 GHz band and the 26MHz within the 900 MHz band. This combination of greater availablebandwidth and reduced sources of interference makes the 5 GHz bands adesirable frequency range in which 4G WLANs have performance comparableto that achieved by wired networks.

Unfortunately, multipath interference, or Rayleigh fading, occurs whenradio waves reflect off the surface of physical objects, producing acomplex pattern of interfering waves which causes signals to meet at theantenna and cancel each other out. In addition, further interference isgenerated by other devices, which operate in the 2.4 GHz frequencyrange. These forms of interference present a design challenge forconventional WLAN systems, which have resulted in many conventional 3GWLANs relying upon a spread spectrum scheme. Unfortunately, byovercoming this interference with spread spectrum technology, such asFrequency Hopping Spread Spectrum (“FHSS”) and Direct Sequence SpreadSpectrum (“DSSS”), these 4G WLANs are limited to data rates of up to 1Mbps to 2 Mbps.

While FHSS and DSSS at first, may appear to be simpler to implement thanother schemes, there still are some very subtle difficulties that occurwhen strong interfering signals exist in such a system. For example, thebasis of the noise immunity within a DSSS-based system is the fact thatthe desired signal and interference (or noise) are uncorrelated. Incomplex interference environments, which are becoming more common asusage increases, particularly ones in which very strong signals may bepresent, non-linearities in the receiver generate intermodulation (“IM”)distortion products between the desired signal and the interferingsignals. These IM products now are correlated with the desired signal,thus reducing the resulting signal to noise ratio when processed in thereceiver. Therefore, even though the use of spread spectrum techniquescombined with more available bandwidth and more complex modulationschemes allows such WLANs to operate at higher data rates, those datarates are not as high as desired (e.g. over 10 Mbps).

Therefore, there currently is a need for a WLAN system, which canachieve higher data rates while still maintaining relative immunity tointerference, such as multipath propagation interference.

DISCLOSURE OF INVENTION

Accordingly, the present invention provides such a need by utilizing afrequency offset differential pulse position modulation scheme totransmit data between computing devices within a wireless network system(100). The differential pulse position modulation component of thescheme enables the present invention to provide relative immunity tointerference for the system (100). In particular, such immunity frominterference is achieved by utilizing a blanking time between pulsepositions, which is large enough to allow the interference betweenpulses to subside and not interfere with the quality of the datatransmission. The frequency offset component of the scheme enables thesystem to utilize multiple frequency channels to enable the system (100)to achieve higher data rates. In particular, by utilizing a time offsetbetween the frequency channels, between which the system switches, theblanking time can be reduced, thereby increasing the amount of data thatcan be transmitted within a set period of time in the system (100).

BRIEF DESCRIPTION OF THE DRAWINGS

These and other more detailed and specific objects and features of thepresent invention are more fully disclosed in the followingspecification, reference being had to the accompanying drawings, inwhich:

FIG. 1 is an illustration of an overview of a block diagram of awireless system of an embodiment of the present invention.

FIG. 2 is an illustration of an overview of a block diagram of atransceiver of an embodiment of the present invention.

FIG. 3 is an illustration of a time domain view of the transmissionscheme of an embodiment of the present invention.

FIG. 4 is an illustration of a frequency domain view of the transmissionscheme of an embodiment of the present invention.

FIG. 5A is an illustration of a more detailed block diagram of anencoder of an embodiment of the present invention.

FIG. 5B is an illustration of a high level flow diagram of a method forconverting from binary data signals to frequency offset differentialpulse position modulation signals in an embodiment of the presentinvention.

FIG. 5C illustrates a detailed block diagram of a transmit MAC statemachine of an embodiment of the present invention.

FIG. 5D illustrates a wait-for-acknowledge state machine of anembodiment of the present invention.

FIG. 6A is an illustration of a more detailed block diagram of anencoder of an embodiment of the present invention.

FIG. 6B is an illustration of a high level flow diagram of a method ofconverting from frequency offset differential pulse position modulationsignals to binary data signals in an embodiment of the presentinvention.

FIG. 6C illustrates the receive MAC state machine of an embodiment ofthe present invention.

FIG. 6D illustrates the timer processes, such as rx_phy timers andcounters, which determine the AGC stabilization time, the IPG time, andinter-pulse timer of an embodiment of the present invention.

FIG. 6E illustrates the receive diversity state machine of an embodimentof the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

Embodiments of the present invention are now described with reference tothe Figures where like reference numbers indicate identical orfunctionally similar elements and the left most digit(s) of eachreference number corresponds to the Figure in which the reference numberis first used.

FIG. 1 illustrates a WLAN system 100 of an embodiment of the presentinvention. The system 100 includes a computing device 105A, a wirelesstransceiver 107A, a wireless transceiver 107B and a computing device105B. Computing device 105A is coupled to wireless transceiver 107A,which together serve as a node on WLAN system 100. Computing device 105Bis coupled to wireless transceiver 107B and together represent a secondnode on the network system 100. Computing device 105A communicates withcomputing device 105B by utilizing wireless transceiver 107A to transmitradio frequency (“RF”) frequency offset differential pulse positionmodulation (“FO-DPPM”) signals to wireless transceiver 107B, which iscoupled to computing device 105B. In one embodiment of the presentinvention, computing devices 105A and 105B can be an Intel-based (e.g.Pentium III) personal computers or handheld computing devices, such as aPalm Pilot-type devices, from such companies as Palm Computing (e.g.,Palm Pilot), Handspring (e.g., Visor) and Microsoft (e.g., Windows CEdevices). In addition, transceiver 107A communicates directly withtransceiver 107B or indirectly via a wireless network router (e.g.,Ethernet router with wireless FO-DPPM capability). One skilled in theart, however, will recognize that system 100 can be modified in numerousadditional configurations to maximize the use of the FO-DPPM schemewithin system 100. For example, one skilled in the art will recognizethat FO-DPPM can be used with any type of wireless scheme including withdifferent frequencies as well as with different network protocols. Inaddition, the computing devices 105A and 105B can be any type ofelectronic device, which can manage and/or display electronic data.Furthermore, the number of nodes, which include a computing device 105and a transceiver 107, is not limited.

The wireless RF scheme of an embodiment of the present inventionimplements the network issues that are specified by the IEEE 802.3standard that are not collision related. The wireless protocol alsoborrows several concepts from the IEEE 802.11 standard. The three phasesof the RF wireless MAC are CSMA/CA/ACK phases. The CSMA phase as withthe Ethernet, each node in the network 100 monitors the RF mediaconstantly for activity prior to commencing its own transmission. A nodedoes not attempt to transmit when the media is sensed to be busy. Inaddition, each node in the network 100 is assessing when the media waslast released, to determine when Inter-packet Gap (“IPG”) expires.

In the CSMA phase, the MAC protocol takes two actions depending uponwhether or not carrier is present when a frame is available fortransmit. In the event media is busy when a frame is ready fortransmission, the node defers until the on-going transmission finishes.Once the ongoing transmission finishes, all nodes with traffic to sendcompute the time of their next transmission attempt by waiting the IPGtime plus a randomized interval (e.g., a random multiple of access-slottime). Should the media become busy again before the node's scheduledtransmit time, a new transmit schedule is computed based upon when thenew transmission finishes plus the another randomized interval. Thesubsequent randomized interval are not correlated in any manner,however, as this process repeats the upper bound for the randomizationinterval grows. If the media is unoccupied when a new frame is availablefor transmit, a node waits out the IPG if any and then performs therandomization process.

In the CA phase, the MAC protocol waits the randomized interval. Therandomized interval is based on the units of access-slot time.Access-slot time can be defined such that all nodes in the network 100can see each other's traffic within this unit of time. Therefore,randomized multiple of this time unit will differ in a such a way thatunless nodes pick the same number of time units, they can see eachother's traffic before attempting to transmit and causing collision. Theaccess-slot time also can be considered the collision window in Ethernetterminology. Elements of the access-slot time to consider are 1)transmit turn-on time; 2) medium propagation time and 3) carrier busydetect. The MAC protocol besides randomizing the access to the mediaalso deploys a linear back off algorithm in dynamically expanding therandomization window based on the media availability or lack thereof.Note that this linear back off is different from the retransmit back-offalgorithm explained below.

In the ACK phase, the MAC protocol waits to receive the acknowledgementpacket once the IPG time elapses. The node that sends theacknowledgement packet will not try the randomization process, whichguarantees that, acknowledge packet is sent unless a hidden node doesnot hear the carrier sense and violates the CA protocol. In the ACKphase, different timeouts are employed to make sure a retransmission canbe scheduled once the acknowledgement is not received. The MAC protocolretransmission can follow the IEEE 802.3 back-off algorithm. The slottime used in the back-off algorithm is a network level quantum of timeand is called retransmit-slot time.

FIG. 2 illustrates a transceiver 107 of an embodiment of the presentinvention. Transceiver 107 includes a transmitter 220, an encoder 230, areceiver 225, a decoder 235, a clock 250, an interface 240 and anantenna array 210. Computing device 105 is coupled to interface 140,thereby allowing one computing device (e.g., 105A) to communicate withanother computing devices (e.g., 105B) within the system 100.

The antenna array 210, which is coupled to transmitter 220 and toreceiver 225, receives FO-DPPM signals from another antenna array. Inone embodiment, the antenna array 210 is a dual antenna configurationand receives single narrow-band frequency (e.g. 5 GHz frequency band)signals. By utilizing a dual antenna configuration, antenna array 210has spatial diversity between the antennas, which reduces noise andmultipath interference that exist within system 100. For example, if oneantenna of the antenna array 210 is experiencing fading, the transceiver107 automatically can switch the receiving of the RF signals to theother antenna within the array 210. Because such high frequencies (e.g.5.8 GHz frequency) have a short wavelength, these two antennas canreside within one radio antenna assembly without significantlyincreasing the size of the overall transceiver 107. When antenna array210 is used at low power consumptions (e.g. using 500 milliwatts or lessof power), the antenna array 210 also complies with the low power rulesof the Federal Communication Commission (“FCC”) Part 15 (unlicensed),subpart 249. Receiver 225 is coupled to antenna array 210 to receiveFO-DPPM signals from the antenna array 110. Receiver 225 utilizes anenvelope detection scheme to convert the FO-DPPM signal from RF signals,which are received by the antenna array 210, to electrical signals. Bymerely using the amplitude of the signal to encode the informationwithin the FO-DPPM data stream, no information is encoded in thefrequency or phase of the signal. Therefore, minimal requirements existfor maintaining phase noise and frequency stability in the receiver 225to detect the FO-DPPM signals. Rather, the receiver 225 merely needs tobe sufficiently stable to be able to detect the desired pulsed signalswithin the specific passband frequency channels.

Transmitter 220 is coupled to antenna array 21 0 and converts FO-DPPMsignals from electrical to RF signals, which then are transmitted viaantenna array 210 for propagation over the system 100. Transmitter 220also utilizes the envelope detection scheme to encode the FO-DPPMsignals, which are received from the encoder 230, to convert theelectrical signals into RF signals. By transmitting the signal basedupon the amplitude of the signal, no information needs to be encoded inthe frequency or phase of the signal, thereby minimizing therequirements for phase noise and frequency stability. The frequencies ofthe transmitter 220, therefore, merely need to be sufficiently stable totransmit the desired FO-DPPM signal within the proper passband frequencychannel.

The receiver 225 and transmitter 220 also utilize a carrier-senseprotocol, similar to Ethernet (IEEE 802.3), to ensure that the FO-DPPMsignals are effectively transmitted and received. When the carrier-sensemechanism determines that the transmission (e.g., RF) medium is busy,the transmitter 220 halts for a short, random back-off period after themedium becomes available before attempting to resend. A conventionalaccess protocol (e.g., IEEE 802.3) or back-off algorithm (e.g.,Ethernet) can be used to ensure that all nodes have access to all othernodes on the system 100. The data is transmitted in frames or packets ina similar manner to that of Ethernet. Each data packet includes aheader, which prepends a standard Ethernet packet, to allow the packetto be transmitted via RF signals. In addition, special maintenancepackets can be transmitted to permit dynamic configuration and controlof the system 100. Furthermore, an acknowledgment-based protocol alsocan be implemented to ensure a reliable link transmission. To enablecompatibility between different versions of FO-DPPM as well asvariations in the transmission protocols, receiver 225 analyzes theprotocol version frame of the preamble of the first data packet of adata stream. Based upon this version information contained within theversion frame of the preamble of the first data packet, receiver 225 canadjust such aspects of its configuration, such as increasing ordecreasing the number of frequency channels that the FO-DPPM signalswill be encoded upon.

The encoder 230, which is coupled to interface 240, transmitter 220 andclock 250, receives binary data, which is received via interface 240from computing device 105, and converts and transmits FO-DPPM signalsvia transmitter 220 to antenna array 210. The conversion of the binarydata to FO-DPPM signals will be discussed in more detail with regard toFIGS. 5A and 5B. The decoder 235, which is coupled to interface 240,receiver 225 and clock 250, receives electrical FO-DPPM signals fromreceiver 225 and converts the FO-DPPM signals to binary data, which aretransmitted via interface 240 to computing device 105. The conversion ofthe FO-DPPM signals to binary data will be discussed in more detail inFIGS. 6A and 6B.

FIGS. 3 and 4 respectively illustrate a time domain view and frequencydomain view of amplitudes in an embodiment of a two-channel (e.g.,channel 1 and channel 2) four (“K=4”) time difference (“pulse position”)FO-DPPM scheme for system 100 of the present invention. One skilled inthe art will recognize that FIGS. 3 and 4 merely are intended toillustrate the principles behind FO-DPPM and are not intended to limitthe scope of the present invention to this one embodiment. One skilledin the art will recognize that the principles discussed in the four timedifference FO-DPPM scheme also will apply to any FO-DPPM schemeincluding those with a smaller or larger number of time differences orFO-DPPM channels.

The time difference between the pulses in each channel of the FO-DPPMdata stream typically is the sum of two segments: 1) a “blanking” timeinterval (“T_(b)”) and 2) a “coding” time interval (“T_(c)”). Theblanking time interval is used to minimize the sensitivity of the system100 to interference (e.g., multipath propagation interference) byproviding enough time between pulse signals to ensure that theinterference has decreased enough to not affect the detection of thepulse position of the FO-DPPM signal. If the blanking time interval istoo short, the interference within the signal will not be effectivelyminimized. If the blanking time interval is too long, bandwidth will gounused, thereby reducing the bandwidth efficiency of the system 100. Thecoding interval is used to encode a specific data value based upon timeposition of the FO-DPPM pulse. For example, depending upon the bitpattern, which the FO-DPPM pulse is intended to represent, the FO-DPPMpulse will be positioned at a specific time position within the set ofpossible time positions. Note that the “differential pulse positionmodulation” aspect of this scheme refers to the fact that theinformation is encoded only in the time difference between successivepulses. As such, an absolute time reference is not needed between anycommunicating nodes.

For the number (“K”) of possible pulse positions (“symbols”), the time(“T_(ensemble)”) to transmit the complete ensemble of all possiblesymbols is(T_(b)+0*T_(c))+(T_(b)+1*T_(c))+(T_(b)+2*T_(c))+(T_(b)+(K−1)*T_(c))=KT_(b)+(K)(K−1) T_(c)/2. Since all symbols are considered to be equallylikely, the average time to transmit one symbol is the total timedivided by the number (“K”) of symbols. Thus, the average time totransmit one symbol is T_(ensemble)/K=(K T_(b)+((K)(K−1) T_(c)/2))/K;and the symbol rate (“Rs”) in symbols/second is R_(s)=K/(KT_(b)+((K)(K−1) T_(c)/2)). Since one embodiment of the present inventionis based upon a binary encoding scheme, the number of bits per symbol isLog₂ (K) bits per symbol. Thus, in a four pulse position (K=4) 4-DPPMchannel, each pulse can represent a 2 bit symbol. In an eight pulseposition (K=8) 8-DPPM channel, each pulse can represent a 3 bit symbol.Therefore, the average data rate (“R_(b)”) in bits per second (“bps”),is the symbol rate (“R_(s)”) multiplied by the bits per symbol (Log2(K)). Thus, the average data rate (R_(b)) of a channel employing theK-DPPM scheme is K Log₂ (K)/(K T_(b)+((K)(K−1)T_(c)/2)).

By implementing FO-DPPM scheme, which includes two frequency offset4-DPPM channels, the data stream is modulated, such that each alternatepulse is transmitted on a different frequency channel, which isseparated by frequency offset (“dF”) and a time offset (“dT”). With twofrequency channels, a time offset (“dT”) is approximately equal to onehalf of T_(b), thus the blanking time T_(b) for each channel iseffectively doubled in length of time. Therefore, the FO-DPPM schemeallows the blanking time T_(b) to be reduced by half while stillmaintaining the same effective blanking time between DPPM pulses in thesame channel. Thus, if a 150 ns blanking time T_(b) was used for asingle channel DPPM scheme to effectively dissipate the multipathpropagation interference, FO-DPPM can effectively reduce T_(b) to 75 ns.In an alternative embodiment, if an adaptive equalizer were be includedwithin the receiver 225 to learn the nature of the multipathinterference within the system 100 and compensate accordingly byadjusting T_(b) on a dynamic basis, the blanking time T_(b) could beeven further reduced. For example, if at any moment during operation ofthe system 100, multipath interference was weaker, a smaller T_(b) couldbe set to further increase the data rate of the system 100. In thisalternative embodiment, blanking time T_(b) could be reduced at times toas little as 50 ns to 25 ns.

In such a two channel 4-DPPM FO-DPPM system 100, the data rate isapproximately 20 Mbs. However, by utilizing a higher order frequencyoffset format (e.g., 3 or 4 frequency 4-DPPM channels), the blankingtime T_(b) can be reduced by a factor of three or four from its originalvalue, thereby increasing the achievable data rate even further (e.g. 30or 40 Mbs). Further, if higher orders of pulse position are implemented,for instance 8-DPPM, in conjunction with higher orders of frequencyoffset, even higher data rates can be achieved. For example, if an8-DPPM, 4 Frequency Offset system were implemented a 60 Megabit/seconddata rate would result.

FIGS. 5A and 5B illustrates a more detailed block diagram of an encoder230 and a flowchart of the conversion of binary data into FO-DPPMsignals, respectively, of an embodiment of the present invention. Toenable compatibility between different versions of FO-DPPM, a protocolversion frame is appended onto the preamble of the first FO-DPPM datapacket of each data stream. This protocol version frame allows thetransmitter 220 of the computing device 105 to communicate to thereceiver 225 of the receiving computing device 105 information, such asthe number of frequency channels, which is being used in the currentversion of the FO-DPPM scheme that is being used.

Encoder 230 includes a transmit MAC (“tx_mac”) module 510 and a transmitPHY (“tx_phy”) module 520. The tx_mac module 510 implements the 4-DPPMprotocol of an embodiment of the present invention. The tx_mac module520 further provides programmability for network parameters. The tx_macmodule 520 also implements the 3 phases of the data transmission,CSMA/CA/ACK. The tx_phy module 520 performs the 4-DPPM encoding and canbe viewed as a slave to the tx_mac module 510 and the receive PHY(“rx_phy”) module parameters.

The tx_mac module 510 includes:

Timer process which implements various system timers.

The transmit MAC state machine which implements the CSMA and CA phase ofthe transmission

The Wait-for-Acknowledge state machine which implements the ACK phase ofthe transmission

The ACK generation state machine which is embodied into the transmit MACstate machine

Transmit Antenna diversity process

CRC-32 generation block (per IEEE 802.3)

In addition, the tx_mac module 510 further includes the followingvarious timers, which implement different components of the RF MAC:

Access-slot time timer. The concept of the access-slot time is based onthe round trip delay between any sending and receiving pairs. This is ameasurable unit in time based of the RF propagation delay and isprogrammable as the distance between the two unit increases. In amulti-node network this value has to be at the minimum the round tripdelay between two sending and receiving pairs that are the farthestapart. This unit of time is used in randomizing the access to theairspace once the airspace becomes free. Since the nodes in the networkare assumed to pick different number of time slots from the randomnumber generator, they will hear each other within one time slot and thenode who had the lowest number will force all other nodes to defer. Theonly chance for collision is if two nodes pick the same number from therandom number generator. Access-slot value is programmable. This valuegets loaded on the access-slot timer every time a slot needs to betimed.

Deferral counter. This is the count of the number of times thetransmit-process defers before accessing the airspace. The maximum valuefor this counter is programmable.

Access-slot window counter. This window determines the maximum number ofaccessslots that a node can pick from the random number generator. Thiswindow is dynamically enlarged if the deferral counter grows. Thestarting value for this counter is programmable.

Retransmit counter. This counter keeps the count of retransmission.Packets are scheduled for retransmission every time an acknowledge frameis not received.

Free running random number generator.

Preamble counter. The number of preamble bytes is programmable. There isa one-to-one correspondence between this value and the programmablereceive diversity preamble count window.

FIG. 5C illustrates a detailed block diagram of a transmit MAC statemachine of an embodiment of the present invention. A transition fromIDLE is cased by the transmit buffer manager having data in its transmitqueue and the airspace being available (as indicated by the rx_phyRX_IDLE signal). Here is a brief description of each state:

TX_RDY state: This is the state from which the transmit-process islaunched. In this state a random number is drawn to see how manyaccess-slot times the transmit-process has to wait (defer) beforelaunching. The random number generation is used to implement thecollision avoidance (CA) phase, which takes place in this state. Atransition is made to TX_DEFER state if the number that is drawn is anon-zero number (which typically is the case since the number zero isreserved for the acknowledge packet). Also every time the deferral takesplace a counter is incremented. The value of this counter is presentedto the transmit buffer manager upon completion of the transit-process.The value of this counter also determines the largest number thetransmit-process can draw from the random number generator. The value ofthis counter increases on average on a per packet transmission as thenumber of nodes in the network increases and thus causing an expansionof the randomization window on a per packet basis. Therefore, thedeferral count is a good dynamic indicator of how busy the network isand as such the wait time for each node before accessing the air growsto minimize the collision probability.

If the deferral counter reaches a programmable maximum, thetransmit-process no longer tries to access the network and exits thetransmit-process. In this case the excess deferral bit in the transmitstatus filed also is set. If TX_RDY state is entered from other states,a transition is made to NW_BUSY state if the airspace is found to bebusy. The transitions between the states TX_RDY, TX_DEFER, and NW_BUSYimplement the CA phase of the wireless MAC.

TX_DEFER state: This is the state in which the transmit-process waitsuntil its drawn deferral access-slots transpires. Following this state,if the airspace is not busy a transmission is launched by moving to theTX_PAMBL state. However if the airspace is found busy a transition toNW_BUSY is made.

NW_BUSY state: This state is reached from TX_DEFER or TX_RDY when theairspace is found to be busy. This state is also entered if a latereception is sensed as transmission is launched.

TX_PAMBL state: In this state the transmission is launched. The preamblecharacter is transmitted a programmed number of times. As this state isentered, the tx_phy kickoff signal (TXENB) is asserted.

TX_SFD state: In this state, one byte of SFD is transmitted.

TX_D state: In this state, the payload is read from the transmit buffermanager and is transmitted.

TX_D_ACK state: This is a parallel state to TX_D state. The payload inthis state is fixed and generated by the tx_mac process. The source anddestination addresses in the acknowledge packet is popped from thereceive MAC address stack in this state and sent along with the fixedheader and the CRC-32 value.

TX_CRC state: In this state the 4 bytes of the CRC-32 value that iscomputed for the payload is transmitted.

WAIT_ACK state: In this state wait for acknowledge state machine iskicked off if the transmitted packet requires an acknowledge. This stateis bypassed if the transmitted packet does not require an acknowledge orthe packet is a multicast packet.

TX_COLL state: This state is reached if an acknowledge is not received.In this state a random number is drawn from the random number generatorto determine how many collision-slot times the transmit-process has towait before attempting to retransmit the packet. The collision-slottimes are fixed to 50 μs. This number is chosen to be very close to IEEE802.3 collision slot time. Multicast packets are sent twice by the txmac 510. This state also is entered if it is the first pass of themulticast packet transmission.

TX_BKOFF state: This is the actual wait state for the retransmitprocess. The transmit-process waits in this state the drawn number ofcollision-slot times before advancing to RX_RDY state. If during thiswaiting period, a request for an acknowledge transmission is received,this state is exited and an acknowledge packet transmission isscheduled. Once the acknowledge transmission is over, the TX_RDY stateis entered with the previous counter and parameter values restored tothe same values in the TX_BKOFF state.

TX_ACK_WAIT_IDLE state: This state is entered when an acknowledge needsto be transmitted. The transmit-process waits for the airspace to becomeavailable a fixed period of time in this state. The request foracknowledge transmission is received at the end of the reception processand as the IPG period is reached. Since the slot zero after the IPG isreserved for acknowledge transmission, a timeout counter is kicked offin this state. The airspace has to become free before this timeoutcounter expires (10 μs), which is usually the case. However if anotherparty violates the IPG and starts transmitting, it is a moot point tosend the acknowledge packet and cause collision. It also is unnecessaryto transmit the acknowledge packet an arbitrary time into the futuresince the original sender waits for acknowledge to be returned for afixed programmable period of times. If the packet is not received withina certain period of time it is assumed that either the acknowledge islost or the responding party did not receive the original message (seewait-for-acknowledge state machine).

TX_DONE state: This is the state that issues the transmit status vectorto the transmit buffer manager. Following this state the state IDLE isreached via the TXCRS_OFF state.

TXCRS_OFF state: This is a dummy state that is there to make sure TX_CRSis de-asserted before transmit-process moves to IDLE state. Since tx_phymodule 520 is behind the tx_mac module 510 by almost a byte due totx_phy and tx_mac pipelining, this state is needed to absorb thepipeline delay. Once the tx_mac module 510 moves to IDLE state, tx_phymodule 520 also is free.

FIG. 5D illustrates a wait-for-acknowledge state machine of anembodiment of the present invention. The acknowledge handshake betweenthe two communicating parties takes place within a specific time of theend of the transmission/reception. In the transmit acknowledge directionthe packet is sent out right after the IPG which will reach thecommunicating party no later than an access-slot time later. The waitingparty for an acknowledgement waits a certain time (see the figure below)from the end of the IPG. Therefore, in the acknowledge transmitdirection time if the airspace does not become available for 10 μs afterthe IPG, the acknowledge transmission is abandoned. In the acknowledgereceive direction as the state machine below indicates, retransmissionis scheduled if acknowledge packet is not received within a certain timeframe.

WAIT_FOR_IPG state: This state is entered upon the end of transmissionand is exited upon assertion of the IPG signal.

IPG state: This state is maintained until the IPG time is over.

WAIT_FOR_CRS: This state is entered upon negation of the IPG signal andis exited if CRS is not received within a time frame that is 2 times anaccess-slot time.

WAIT_FOR_ACK state: This state is entered upon reception of the RX_CRSsignal and is exited upon either reception of the ACK_RCVD signal fromrx_mac or a timeout from the ack-packet-length timer. Theack-packet-length timer corresponds to the time it takes to transmit thelongest preamble byte count plus the 20-byte acknowledge packet. Thistime is measured from the assertion of the RX_CRS. ACK_RCVD should beasserted during this time period.

Transmit Diversity process. The transmit diversity process determinesthe antenna. The antenna value parks on the last transmit antenna value.During the receive, the rx_phy module 610 can toggle this value throughthe receive diversity state machine. Every time the antenna is switchedan AGC_DIS is asserted to the Radio and the subsequent pulses are maskedoff by the rx_phy module 610 for a period of 2.775 μs (2.5 μs per radiospecification+one symbol time). Here is the step by step choice for theantenna value Upon transmission kickoff the value programmed by thesoftware is driven onto the antenna.

If the transmission is for an acknowledge packet, the value receivedfrom rx_phy module 610 is driven onto the antenna. Note that as soon asthe reception is over, the antenna switches back to its previoustransmission state, therefore for acknowledge packet transmission, thevalue remembered in the rx_phy module 610 and passed on to the tx_mac520. Antenna switch if it happens at the end of the reception for whichan acknowledge needs to be sent is unnecessary. Future transmitdiversity algorithms should remedy this. Antenna switch if it happensbetween back to back reception with or without acknowledge is alsounnecessary. Future transmit diversity algorithms should remedy this.

Every time a retransmit is scheduled the antenna value is toggled fromits previous value.

For broadcast packets the antenna selected is forced to the value ‘0’for the first transmit pass and to the value ‘1’ for the second transmitpass.

Transmit Phy (“tx_phy”) module 520 includes the transmit encoder and theAGC block. The transmit encoder process loads a byte from tx_mac module510 onto a parallel to serial converter. The byte is examined 2-bit at atime and a down counter is preloaded with the following values:

12 for the Symbol 00

15 for the Symbol 11

18 for the Symbol 01

21 for the Symbol 10

The counter then runs until it reaches 0 at which time a pulse isgenerated. This process continues until all bytes are transmitted. Thetx_phy module 520 makes no assumption about the preamble value, thestart-of-frame delimiter or the payload type. It simply performs 4-DPPMencoding on the bytes received from tx_mac module 510.

The AGC block primary function is to assert the AGCDIS_signal to theRadio. It asserts AGCDIS_(1')upon detecting one of the followingconditions. This signal is pulsed for approximately 300 ns and driven toradio in the following cases:

At the end of receive process (RX_EOF)

At the end of the transmit process (TX_EOF)

When antenna switches

The AGC block also drives the antenna value to the Radio. The antennavalue is the tx_mac value except for duration of RX_CRS. During RX_CRSrx_phy can switch antenna value by asserting TOGGLE_ANT which is drivenfrom its receive diversity state machine.

FIGS. 6A and 6B illustrate a more detailed block diagram of the decoder235 and a flowchart of the conversion of FO-DPPM signals into binarydata, respectively, of an embodiment of the present invention.

Decoder 235 includes a receive PHY (“rx_phy”) module 610 and a receiveMAC (“rx_mac”) module 620. The parameters of the receive PHY (“rx_phy”)module 610 are coupled with the radio behavior. The receive MAC(“rx_mac”) module 620 converts serial data to parallel and stores theIEEE 802.3 source and destination addresses for acknowledge packettransmission.

Receive MAC (“rx_mac”) module 620 receives serial data from the rx_phymodule 610 and converts the serial data into parallel data beforesending the data to receive buffer manager. Receive MAC also stores theEthernet source and destination address fields for the acknowledgetransmission. The address filtering is performed in the receive buffermanager. rx_mac module 620, however, detects that a received packetneeds an acknowledgement reply and sends the request for acknowledgetransmission to the tx_mac module 510.

FIG. 6C illustrates the receive MAC state machine of an embodiment ofthe present invention. The receive MAC state machine is initiated whenstart-of-frame delimiter is detected. The state machine is exited whenthe last indication is received from the rx_phy module 610 when thereceived packet does not need an acknowledge reply or when theacknowledge packet has been transmitted. No specific actions take placeanytime during any state. In addition to the state machine reportingstatus on the signal rx_busy, the state machine also generates theack_req in the state IPG.

Receive PHY (“rx_phy”) module 610 includes the diversity state machineand numerous counters to assist in timing different relative events. Theprincipal basis of rx_phy operation is the detection of the CarrierSense (RX_CRS). Unlike the wire-line detection of RX_CRS, which is thepresence of energy on the wire, RX_CRS is detected through sampling ofreceived pulses from the Radio. Several timers and counters are includewithin the rx_phy module 610 to measure the inter-pulse periods forsymbol decoding and various symbol arrival order and arrival timeintervals. Through these measurements the Carrier Sense is detected. Asignal that is received by the rx_phy module 610 may be quite distortedfrom its original from due to multi-path and fading effects. To have arobust reception, rx_phy module 610 chooses the correct antenna, whichprovides the best reception, and make a few assumptions about the natureof received traffic. Further it has to deal with the nodes that areoutside of the Carrier sense domain and do not necessarily synchronizeto other nodes transmission. rx_phy module 610 has to be aware of theRadio parameters such as AGC stabilization time to mask off receptionduring AGC stabilization periods. Therefore, the timer and counterswithin rx_phy module 610 are utilizes to assist in reception orrejection of received frames. The receive diversity algorithm also isdependent on AGC stabilization time since upon antenna switch, thereceive diversity state machine will mask off reception until the AGCstabilization period is over.

FIG. 6D illustrates the timer processes, such as rx_phy timers andcounters, which determine the AGC stabilization time, the IPG time, andinter-pulse timer of an embodiment of the present invention. Thefollowing is the list of some of the timers that are included within therx_phy module 610.

IPG timer, (which is a 5 μs from the end of TX_CRS or 4.125 μs from theend reception).

Inter-pulse timer, which determines the symbol type. The inter-pulsetimer counts the number of 80 MHz clock ticks and based on the countdetermines the symbol type or a symbol error. Symbol error is indicatedif no pulse is received within 275 ns of the last received pulse. When apulse is detected and the inter-symbol timer starts, all incoming pulsesare masked of for the next 10 clock ticks (125 ns) and the 4-ppmdecoding starts after 125 ns has elapsed from the first detected edge.

Symbol 00 range (137.5-162.5 ns, 11-13 clocks).

Symbol 11 range (175-200 ns, 14-16 clocks).

Symbol 01 range (212.5-237.5 ns, 17-19 clocks).

Symbol 10 range (250-275 ns, 20-22 clocks).

Every time a pulse is detected, the symbol decode clock counter isreset, thus input jitter and wander are not built up. The jitter allowedon reception of each symbol is +/−12.5 ns.

End of reception timer. This is an 825 ns period from the last pulsereception in which no pulse is received. End of reception is indicatedupon the timer reaching this value.

HOLD_OFF signal assertion timer. This timer asserts HOLD_OFF signal 1.8μs after reception of the first pulse.

AGC stabilization period timer. This timer is activated upon detectionof AGCDIS_signal. When this signal is asserted the data received duringthe AGC stabilization time (maximum of 2.5 μs per Radio specification)can be ignored. rx_phy module 610 can mask the incoming data for 2.775μs which is 2.5 μs plus 1 symbol time (maximum of 275 ns). Following AGCstabilization period, a pulse is received in 287.5 ns for receivediversity to start preamble decoding. If no pulse is received in 287.5ns, a symbol error is indicated and antenna switching takes place.Following the 1^(st) antenna switch, another AGC stabilization period isobserved after which the data is being received and no further antennaswitch takes place. No more than one TOGGLE_ANT is issued to the AGCblock (see antenna diversity state machine for more details).

Receive diversity sample size down counter. This counter is preloadedwith the programmed value. If upon reaching the count of Zero no symbolerror has been detected, the receive diversity state machine does notswitch antenna. Every time the airspace is not busy and subsequently apulse is received this counter gets preloaded with the programmed value.

TX_CRS extension timer. This timer extends the TX_CRS by 300 ns.

Pulse counter during IPG. This counter counts consecutive receivedpulses during the IPG. 4 consecutive pulses during IPG asserts theRX_CRSP which in turn causes adjustment to the IPG. These consecutivepulses are considered to be valid reception and as a result, the IPG ismeasured from the end of this newly asserted RX_CRSP (825 ns+4.125 s).The sole reason for the assertion of RX_CRSP and existence of RX_CRSPplus is to adjust the IPG in case of a fade-out. In all other casesRX_CRS and RX_CRSP typically are the same.

FIG. 6E illustrates the receive diversity state machine of an embodimentof the present invention. The receive diversity state machine followsthe expected sequence of 00,11,01,10 symbols (from left to right). Ifthis sequence is not received within the diversity sample size window,antenna is switched and an AGC attack time is waited before receivingfurther symbols. Note that Start of frame delimiter symbol is00,00,10,11 (from left to right).

To further illustrate the two frequency channel FO-DPPM scheme of anembodiment of the present invention, the following example of thetransmission and receipt of FO-DPPM signals between computing devices105A and 105B now will be discussed. Computing device 105A initiates thetransmission of data to computing device 105B by transmitting binarydata from computing device 10SA to interface 240A of transceiver 107A.Encoder 230A receives the binary data from computer 105A via interface240A. The encoder 230A transmits the electrical version of FO-DPPMsignal to transmitter 220A where the transmitter converts the signal toan RF format and transmits with the antenna array 210 the first pulse ofthe FO-DPPM signal on channel 1. The transmitter 220A then waits apredetermined interval, which is less than T_(b) and transmits onantenna array 210 a second FO-DPPM pulse on channel 2. Aftertransmitting the pulse on channel 2, transmitter 220A switches back tochannel 1 and transmits, after a predetermined interval of time, thenext FO-DPPM pulse of the current data stream. The transmitter 220Acontinues to switch back and forth consecutively between the twochannels until the transmission of the data stream is complete. Separatetime interval clocks are maintained for each channel to decode the twoseparate symbol streams. By multiplexing the transmitted data streamsuch that odd numbered symbols are transmitted on one channel (e.g.,channel 1) and even numbered symbols are transmitted on the otherchannel (e.g., channel 2), the effective data throughput can beincreased by the number of channels employed. For example, by utilizingtwo channels, the throughput of the system 100 is double that of asingle frequency channel FO-DPPM scheme.

Receiver 225B receives the FO-DPPM data stream by starting on channel 1and receiving via the antenna array 210B the first FO-DPPM pulse. Thispulse will assist the receiver 225B determine which version of theFO-DPPM scheme the FO-DPPM pulse is related, thereby allowing thereceiver 225B to adjust to the number of frequency channels used in thisversion of the FO-DPPM scheme. Since in our example the FO-DPPM schemeuses only two frequency channels, the receiver 225B will detect thatthis version is using only two channels and, therefore, will begin toalternate between the two frequency channels in order to detect thepulse positions of the FO-DPPM data stream.

The foregoing description of embodiments of the present invention hasbeen presented for purposes of illustration and description. It is notintended to be exhaustive nor to limit the invention to the precise formdisclosed. Many modifications and variations are possible in light ofthe above teaching. For example, the number of frequency channels andtime differences can be increased or decreased depending upon the amountof throughput needed for the system 100. In addition, various protocolscan be used with the FO-DPPM scheme. Embodiments were chosen anddescribed to best explain the principles of the invention and itspractical application to thereby enable others skilled in the art tobest utilize the invention in various embodiments and with variousmodifications as are suited to the particular use contemplated. It isintended that the scope of the invention be defined by the claims andtheir equivalents.

What is claimed is:
 1. A method for transmitting data comprising: afirst frequency channel transmitting a first pulse position modulationsignal; a second frequency channel transmitting a second pulse positionmodulation signal, wherein said transmitting data is alternativelyencoded within the first pulse position modulation signal and the secondpulse modulation signal.